Systems and methods for using cascoded output switch in low voltage high speed laser diode and EAM drivers

ABSTRACT

High frequency laser diode (LD) and electro-absorption modulator (EAM) integrated circuit drivers using a cascaded output switch architecture that increases the output current and voltage edge speed and reduces the peaking and ringing of the output waveform, thus improving the deterministic jitter performance. Also disclosed is a method and apparatus that provides a modulation current dependence of both turn-on and turn-off driving currents that lead to an optimal compromise between the edge speed and output overshoot for a wide range of modulation currents. A PTAT temperature dependence of both voltage swing and current level in the predriver assures a low variation of the overshoot and rise/fall time over a wide temperature range. Using the cascaded output switch architecture provides an easy way of on-chip summation of the modulation and bias currents. Biasing the cascode device with a supply and modulation current dependent base voltage provides an optimum headroom output switch.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to the field of high speed, wide outputcurrent/voltage range laser diode/EAM driver circuits.

2. Prior Art

A laser diode (LD) is an opto-electronic device that provides an outputlight beam when the current through it goes over the lasing thresholdcurrent.

An electro-absorption-modulator (EAM) is an opto-electronic device thatmodulates the intensity of an incoming light beam based on the level ofan electrical control voltage.

A bias current is a constant current that places an opto-electronicdevice just over the lasing threshold.

A modulation current is a switching current that brings theopto-electronic device further into the lasing mode.

A laser diode/EAM driver is a circuit that provides the control currentsand voltages to an opto-electronic lasing device. It consists of acascade of current switching stages and buffers that provide theappropriate amount of voltage and current gain, assuring optimalswitching of the lasing device.

Edge Speed Improvement

The use of laser diodes with high extinction ratios requires laserdrivers that are able to switch large currents. This results in theusage of large output transistors that have high feedback(Cbc—base-collector or Cgd—gate-drain) capacitances. These feedbackcapacitances (Cbc in FIG. 1 and FIG. 2) appear multiplied by the gain ofthe output switch (through the Miller effect—C_(Miller)) at the input ofthe differential output switch, heavily loading the predriver andtherefore significantly reducing the edge speed. The high current in theoutput switch results in a high voltage gain, resulting in a morepronounced Miller effect, and also a higher base-emitter charge storagecapacitance (Cbe=Cpi) that adds to the capacitive loading of thepredriver and further slows the output current/voltage waveform. Thereis a significant speed penalty for EAM drivers (that operate in a 50 Ohmenvironment) and laser drivers working in high impedance environments(Z₀>25–30 Ohms).

Several methods have been used in the prior art to improve the edgespeed. A first solution is to introduce emitter degeneration resistors(Rdegen) as presented in FIGS. 3 a and 3 b for LD and EAM drivers,respectively. The emitter degeneration decreases the voltage gain of theoutput switch and thus reduces the capacitive loading on the predriverdue to the Miller effect. The emitter degeneration also reduces theloading on the predriver due to the lower effective base-emitter (Cpi)capacitance. Reducing both capacitive loadings on the predriver resultsin a significant speed-up of the driving voltage at the input of theoutput switch, and therefore a faster output waveform is obtained.

The foregoing method works well at high supply voltages or at lowmodulation currents where the voltage drop across the degenerationresistor does not significantly impact the headroom available to theswitch. The major drawback of the emitter degeneration technique is thatit requires a high voltage drop across the degeneration resistance.Specifically, to obtain a low voltage gain, the degeneration resistanceneeds to be a good fraction of the load resistance, making the circuitinoperable at high modulation currents and low supply voltages. Inaddition, the layout is not as compact, leading to more metalconnections between transistors and degeneration resistors that addsignificant parasitic capacitances. These extra emitter capacitancesenhance the peaking of the output waveform, requiring more RCcompensation to be used. This slows down the edges, and thus part of thespeed-up advantage given by the emitter degeneration is lost.

A second method used in the prior art to speed-up the driver is to addinductive peaking in the collector (drain) of the output switch aspresented in FIG. 4. However, on-chip inductances capable of passinghigh current levels have high parasitic capacitances that short them athigh frequencies. This is why in most cases the inductive peaking isdone using bond-wire inductances (Lpeak). The major drawback of theinductive peaking is that it trades additional edge speed for moreovershoot of the output waveform that gives supplementary deterministicjitter.

The inductive peaking works fairly well for the EAM drivers that use asymmetric differential output switch and have higher output impedances.In the case of laser drivers that in most cases require an opencollector output switch and usually operate in much lower impedanceenvironments (10–20 Ohm versus 50 Ohm for EAM drivers), adding seriesinductance to speed up the edges gives excessive overshoot that needs tobe damped with additional RC compensation, which in turn slows down theedges. Also if the series inductance becomes high so that the L/R timeconstant becomes comparable with the data rate, the series inductancecan even lead to a slowdown of the edge speed.

In the case of EAM drivers, the inductive peaking is performed byinductances not in the path to the EAM device, and therefore can be wellcontrolled. In the case of laser drivers, the inductive peaking isdifficult to control, as the inductive peaking element is in the pathfrom the laser driver to the laser diode. This path is layout specificand varies from one assembly to another.

A third method used in the prior art to speed-up the driver is theneutralization of the Miller effect. This was done by adding two Millereffect cancellation capacitances (Ccancel) to the differential pair,each from the base of a respective device to the collector of theopposite device (in FIG. 5 from the base of Q1 to the collector of Q2and from the base of Q2 to the collector of Q1). If these two capacitorsclosely match the base-collector capacitance (Cbc) of the output switchtransistors (Q1 and Q2) they can provide a precise cancellation of theMiller effect. In real circuits, there will always be a mismatch thatwill reduce the Miller cancellation effect. Good matching can beachieved by using transistors of the same size as the output switchtransistors for the Miller cancellation capacitances. The major drawbackof this Miller cancellation technique is that the cancellation devicessignificantly increase the output capacitance of the driver and thusreduces the frequency of the ringing that appears during the switchingprocess. If the ringing frequency comes close to the data-rate, itcannot be filtered-out by the SONET filter and will seriously increasethe deterministic jitter. This Miller cancellation technique works wellwhen the predominant slowing down effect is the capacitive loading ofthe predriver and not the output capacitance of the driver.

Temperature Compensation

The prior art has used various temperature compensation techniques toimprove the laser diode switching behavior over temperature. Onetechnique uses a laser driver that gives a modulation current thatincludes a positive temperature coefficient (PTAT) to compensate for theeffects of the laser diode temperature increase.

Another technique uses a temperature dependent current (Itemp) toregulate the common-mode voltage at the bases of the switches Q1 and Q2as shown in FIG. 6. By compensating for the temperature dependence ofthe VBE voltage of the output switch, the collector-emitter voltage(headroom) of the switch is maximized for a given power supply, assuringa higher edge speed.

Overshoot and Rise/Fall Time Control with Modulation Current Dependenceof the Predriver Circuits

Actual laser drivers are required to operate over a wide modulationcurrent range. Optimizing the rise/fall time and theovershoot/undershoot of the output waveform (current for a LD andvoltage for an EAM) requires a modulation current dependence of thepre-driver currents.

Most of the prior art uses standard emitter followers in the pre-driver(Q3 and Q4 in FIG. 7). The drawback of this architecture is that itgives the same value of current for both turn-on and turn-off, leadingto significant overshoot at turn-on when a high edge speed at turn-offis required.

An improvement of the standard emitter follower predriver architectureis presented in FIG. 8. It consists of using a dynamic emitter follower(devices Q3, Q4 and Q7, Q8) that has a different tail current at turn-on(Ief) and turn-off (Ief+Imod/M). This will bring a compromise betweenthe rise/fall time and the overshoot. The drawback of this architectureis that it uses a constant turn-on current for all the modulationcurrent levels, leading to excessive overshoot at low modulationcurrents. Another drawback is that the dynamic emitter followers requirean additional driving voltage Vin* in phase opposition with the maininput voltage Vin. The delay time from the Vin* input to thedifferential pair Q7, Q8 needs to be smaller than the delay time fromthe main input Vin to the Q3, Q4 emitter followers in order that thepull-up and pull-down current levels are set-up correctly. Achievingthis delay time constraint requires a high current consumption in theadditional driving path.

Another method to reduce the output rise/fall time is to use eithersymmetric or asymmetric dynamic coupled emitter followers (Q3, Q4 andQ7, Q8) that injects capacitive charging currents in the output switch,enhancing the peaking and therefore speeding-up the edges, as shown inFIG. 11. The advantage of this method over the traditional inductivepeaking is that the edge speed is improved without worsening the ringingof the output current/voltage (without affecting the damping of theoutput RLC circuit). The drawback of this architecture is the increasedsupply current and large area of on-chip capacitance required for thedynamic coupling of the emitter follower. Furthermore, the overshootcancellation is fixed and does not track with the external elementlayout.

On-Chip Versus Off-Chip Summation of Bias and Modulation Currents

The independent control of the bias and modulation current is achievedin most of the prior art by using two separate current sources. Thesummation of the bias current (Ibias) to the modulation current (Imod)is usually done off-chip by using a high value inductance (Lbias) tominimize the capacitive loading of the driver output by the biascircuitry (Cbias) as shown in FIG. 9.

The direct summation of the bias and modulation current at the driveroutput brings a severe edge speed penalty due to the capacitive loadingof the driver output. One solution to this problem, as presented in FIG.10, is to eliminate the separate bias circuit and to use a differentialoutput pair (Q1, Q2) in which the devices switch between two on-statecurrent levels. This eliminates any additional capacitive loading fromthe bias circuit. This architecture significantly improves the edgespeed due to the switching between two on-state current levels, which ismuch faster than the on-off switching.

Predriver Current Control

Most prior art laser/EAM drivers use off chip control voltages to adjustboth the voltage swing (Imod/N) and the pre-driver current levels(Imod/M) (see FIG. 7). These external adjustments are meant to optimizethe switching performance when operating over wide modulation currentand temperature ranges.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram for a prior art DC coupled laser diodedriver.

FIG. 2 is a circuit diagram for a prior art DC coupled EAM driver.

FIGS. 3 a and 3 b are circuit diagrams for a prior art output switchwith emitter degeneration for use as an LD driver, and for use as an EAMdriver, respectively.

FIG. 4 is a circuit diagram for a prior art output switch with inductivepeaking.

FIG. 5 is a circuit diagram for a prior art output switch with Millercompensation.

FIG. 6 is a circuit diagram for a prior art LD/EAM driver withtemperature compensation of the output switch headroom.

FIG. 7 is a circuit diagram for a prior art LD/EAM driver withmodulation current dependence of the predriver current level and voltageswing.

FIG. 8 is a circuit diagram for a prior art LD/EAM driver with dynamicemitter follower to assure different turn-on and turn-off drivingcurrents.

FIG. 9 is a circuit diagram for a prior art LD/EAM driver with off-chipsummation of the modulation and bias currents using a high valueinductance.

FIG. 10 is a circuit diagram for a prior art LD/EAM driver thateliminates the separate bias current by using a differential pair thatswitches between two on-state current levels.

FIGS. 11 a and 11 b are circuit diagrams for prior art dynamic emitterfollowers used to reduce the output overshoot comprising a balanceddynamic emitter follower, and a one-sided dynamic emitter follower,respectively.

FIG. 12 is a circuit diagram for a cascode output switch LD/EAM driverin accordance with the present invention.

FIG. 12 a is a circuit diagram for an exemplary cascode bias circuitthat may be used in the circuit of FIG. 12.

FIG. 12 b are curves illustrating the Imod, Ibias and VCC dependence ofthe collector emitter voltage of the output switches using the cascodebias circuit of FIG. 12 a in the circuit of FIG. 12.

FIG. 13 is a circuit diagram for an exemplary embodiment of the cascadedoutput switch LD/EAM driver architecture of the present invention.

FIG. 14 is a circuit diagram for another exemplary embodiment of thecascaded output switch LD/EAM driver architecture of the presentinvention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 12 presents the principal architecture of the cascoded outputswitch laser diode/EAM driver of the present invention. This driverassures both fast switching and a relatively low supply voltage VCCoperation, dependent on the required output voltage swing Vswing. As aminimum supply voltage, VCC≈Vswing+1.5V.

The circuit of FIG. 12 has power supply connections V+ (VCC) and V−(circuit ground), control inputs Vmod control and Imod control and apair of complimentary differential input signals Vin and Vin*. DevicesQ10 and Q11 are coupled as a differential pair having a tail currentwith a proportional to absolute temperature (ptat) component Idif_ptatfrom a PTAT Bandgap Reference, and a substantially constant componentIdif_var as set through the Imod control terminal. Resistors Rc act asload resistors for devices Q10 and Q11, with the differential voltagesacross resistors Rc driving the bases of emitter followers Q5 and Q6.Devices Q5 and Q6 have a bias current also having a proportional toabsolute temperature component Ief_ptat and a substantially constantcomponent set externally through the Imod control. Devices Q5 and Q6 mayfurther have an additional bias current, itself having a proportional toabsolute temperature component Ipulldw_ptat and a substantially constantcomponent Ipulldw_var, also set externally through the Imod control,depending on whether device Q7 directing the additional current throughdevice Q5, or device Q8 directing the additional current through deviceQ6, is on. By proper phasing of the input signal Vin* with respect tothe basic input signal Vin, the turn-off current for devices Q5 and Q6may be made larger than the turn-on current for these devices. Thisprovides control of the overshoot and rise/fall times with temperaturethrough the PTAT dependence, and control of the modulation current bythe external setting of the Imod control.

The output of emitter followers Q5 and Q6 is coupled to the bases of theoutput switches Q1 and Q2, coupled as a differential pair with a tailcurrent Imod flowing through resistor Rmod and inductance Lmod,capacitance Cpar being the associated parasitic capacitance. The voltageacross the resistor Rmod is fed back through the negative input ofoperational amplifier OAmod. The positive input of operational amplifierOAmod is externally provided through the Vmod control, as divided downby resistors Rmod1 and Rmod2. The output of the amplifier OAmod iscoupled to the base of transistor Q9, which controls the voltage at thecommon connection of the load resistors Rc. When the differential inputVin is positive, device Q10 will be on and device Q11 will be off. Thusthe base of device Q6 will be at a higher voltage than the base ofdevice Q5, causing the base of switch Q2 to be higher than the base ofswitch Q1, turning switch Q2 on and switch Q1 off. Under theseconditions, the voltage fed back to the negative input of theoperational amplifier OAmod will be equal to the voltage drop acrossresistor Rmod which, neglecting emitter currents, will be equal to thevoltage at the emitter of device Q9 minus the VBEs of devices Q6 and Q2.(The current in the load resistor Rc associated with device Q11 beingzero at this time because of device Q11 being off.) Similarly, in theother half cycle, the voltage across resistor Rmod is equal to thevoltage on the emitter of device Q9 minus the VBEs of devices Q5 and Q1.Thus the Vmod control sets the voltage across resistor Rmod, and thusthe modulation current or tail current Imod for output switches Q1 andQ2, whereas the Imod control sets a component of the tail current fordevices Q10 and Q11, a component of the bias current for devices Q5 andQ6 and a component of the tail current for differential pair Q7 and Q8.

Devices Q3 and Q4 are coupled as cascode devices for the output switchesQ1 and Q2, respectively. The base voltages for the cascode devices Q3and Q4 are set by the output of the operational amplifier OAcasc. Inparticular, the operational amplifier OAcasc drives the bases of cascodedevices Q3 and Q4 so that the emitter voltage for device Q4 is equal tothe output of the Cascode Bias Circuit. Thus the output of the amplifierOAcasc will generally be one VBE above the input voltages thereto. Sincethe emitter voltages of output switches Q1 and Q2 are set through theVmod control, the cascode bias circuit effectively sets the collector toemitter voltages on the output switches Q1 and Q2. Also as may be seenin FIG. 12, a bias current Ibias is coupled to the emitter of cascodedevice Q4 to provide the desired bias current to the EAM/LD.

The cascode bias, together with associated parts of the circuit of FIG.12, is shown in FIG. 12 a. The bias node for the cascode devices,vcascode, is dynamically adjusted as a function of the modulationcurrent I_(mod) and bias current I_(bias), and the power supply voltageVcc. This allows the output voltage to be split across the cascodedevices (Q3 and Q4) and switching transistors (Q1 and Q2) for optimalswitching of the output. Also, by dynamically allocating the outputvoltage across the cascode and switching transistors as a function ofmodulation/bias currents, and power supply voltage, breakdown hazardsare avoided at the output under conditions of low modulation/biascurrents, and high supply voltages. The cascode bias voltage, vascode,is generated by the feedback loop consisting of the output of thecascode bias circuit and amplifier OAcasc which is compared to a voltagewhich includes the base-emitter drop of cascode devices Q3 and Q4. Byincluding the base-emitter drop of the cascode devices in the feedbackloop, temperature and process variations are dynamically adjusted by thefeedback loop, maintaining the desired collector-emitter voltage acrossthe cascode and switching devices.

The cascode bias circuit consists of three current sources, which areproportional to the modulation current Imod, bias current Ibias, andsupply voltage Vcc. These currents are then summed into a loadresistance R_(L), which is connected to the power supply voltage Vcc.Any change in these currents will modulate the voltage drop across theload resistor R_(L). The voltage drop across the load resistor adjuststhe voltage vcascode, which in turn sets the voltage across the cascodedevice Q4 and the switching device Q2.

A typical response of the casode bias circuit is shown in FIG. 12 b. Forincreasing modulation (Imod) or bias (Ibias) current, the voltage dropacross the load resistor R_(L) will increase, causing the voltagevcascode to decrease. This in turn causes the voltage across the cascodeand switching devices to decrease. By appropriately setting the gain forthe modulation and bias current sources, an optimal distribution ofcollector-emitter voltage across the casode device Q4, and switchingdevice Q2 can be obtained. Also, a current proportional to the supplyvoltage Vcc is generated and summed into the load resistor R_(L). Ifthis current were not generated, any increase in the supply voltagewould only increase the voltage across the switching device Q2. Thiscould lead to excessive collector-emitter voltage across the device Q2under conditions of high supply voltage, potentially causing a breakdownhazard. Also, by distributing any increase in the supply voltage acrossthe cascode device and switching device, optimal switching performancecan be obtained at the output.

Now referring to FIG. 13, a circuit similar to FIG. 12 may be seen. InFIG. 13, however, the bias current Ibias for cascode device Q4 isexternally adjustable by a bias control signal applied to the terminalconnected to the resistor Rbias1. In particular, the bias voltage isdivided down by resistors Rbias1 and Rbias2 to provide a positive inputto the operational amplifier Oabias. The amplifier controls the base ofdevice Q15 to provide a current Ibias through the resistor Rbias tocause a voltage drop across the resistor Rbias equal to the divided downbias control voltage applied to the positive input of the amplifierOAbias. Thus the minimum current through the EAM/LD may be setexternally to best match the characteristics of the EAM/LD.

Also in the circuit of FIG. 13, the Vmod control input is used to notonly control the tail current for output switches Q1 and Q2, but alsothe tail current for devices Q10 and Q11 as well as the bias currentsfor devices Q5 and Q6. In particular, the Vmod control input, as divideddown by resistors Rmod1 and Rmod2, is applied to the positive input ofoperational amplifier OAdriver, controlling the bases of devices Q12,Q13 and Q14 to cause a voltage drop across resistor Rdif and resistorsRef equal to the divided down Vmod control voltage. This, then, controlsthe modulation dependent component of the tail current in devices Q10and Q11 and in the bias currents for devices Q5 and Q6. Finally, in thisembodiment, the difference in the turn-on and turn-off currents fordevices Q5 and Q6 is set by the higher voltage of the differential inputVin*, the voltage across the resistor Rpulldw being one VBE below thepositive or the negative side of the differential input, whichever ishigher at the time.

Now referring to FIG. 14, another alternate embodiment of the presentinvention may be seen. This embodiment is similar to the embodiment ofFIG. 13 in many respects, though it generates the equivalent of Vin* ofFIG. 13 as part of the integrated circuit, and also controls theincrease in the turn-off current for devices Q5 and Q6 through the Vmodcontrol input. In particular, the Vmod control input as divided down isalso coupled to the positive input of the operational amplifierOApulldw. The negative input to the operational amplifier OApulldw iscoupled to the pull-down resistor Rpulldw so that the output of theoperational amplifier OApulldw will control device Q22 to control thevoltage at the common connection of the load resistors Rc1. This setsthe voltage across the resistor Rpulldw equal to the voltage on the Vmodcontrol as divided down by the resistors Rmod1 and Rmod2. In particular,the voltage on the common connection of the load resistors Rc1 fordevices Q20 and Q21 will be controlled by the feedback through theoperational amplifier OApulldw to equal the voltage across the resistorRpulldw, plus the VBE of device Q7 or Q8, plus the VBE of device Q16 orQ17, plus the VBE of device Q19 or Q18, depending on which set ofdevices is on at the time. Thus through this closed loop, the Vmodcontrol signal controls not only the tail current for transistors Q10and Q11 and the turn-on current for transistors Q5 and Q6, but also theextra current component Ipulldw_var for the turn-off current for devicesQ5 and Q6.

In the embodiment of FIG. 12, the inductance Lmod and associatedparasitic capacitance Cpar are shown, but are not shown in FIGS. 13 and14. They are equally applicable to FIGS. 13 and 14, however, and are notshown for clarity reasons, to leave drawing space for the additionalother circuitry illustrated in these Figures.

In the present invention, the capacitive loading effect of the outputswitches (devices Q1 and Q2) on the predriver output (emitter followersQ5 and Q6) is significantly reduced by using the cascode devices Q3 andQ4. These devices provide a low impedance at the collectors of theoutput switch and thus minimize the Miller multiplication effect of thebase-collector capacitance of devices Q1 and Q2. As a result, thedriving voltage at the bases of the output switch will have an increasededge speed, leading to faster or sharper edges of the outputcurrent/voltage.

Emitter followers running at high collector current and driving largecapacitive loads lead to excessive voltage peaking that will betransferred into peaking of the output waveform, increasing thedeterministic jitter. Using a cascoded output switch minimizes thecapacitive loading of the last emitter follower of the pre-driver (Q5,Q6) and thus also minimizes the output peaking.

The cascode device from the active side (Q4) has the bias current summedto its emitter. As the summing is done at a low impedance node, thecapacitive loading due to the bias circuit (Cbias) has negligible effecton the output edge speed. Furthermore, the voltage on the bias circuitis kept fairly constant, minimizing the modulation of the bias currentby the output voltage.

This method of bias current summation is particularly suited for EAMdrivers that do not need an additional series damping resistor(Rdamp—see FIG. 12) and that have very wide output voltage swings (ashigh as 3V). It assures an easy way of on-chip summation of the bias andmodulation currents, without any significant penalty on the edge speed,while eliminating the expensive and large ferrite bead used in the priorart off-chip summation.

The cascode device on the dummy side (Q1) also has a permanent currentthrough it, provided by the current source Idummy. This keeps thecascode device in the on-state all the time, significantly reducing itsswitching time by switching between the two on-state current levels,Idummy, when the output device Q1 is off, and Imod+Idummy when theoutput device Q1 is on.

The speed of the switching is critically dependent on the headroom(collector-emitter voltage) of both the switches Q1 and Q2 and thecascode devices Q3 and Q4. The separate Cascode Bias Circuit isimplemented to assure the optimal headroom to the cascoded output switchdevices as the supply voltage (VCC), modulation (Imod) and bias (Ibias)current changes. This circuit was shown in FIG. 12 b. The collectoremitter voltages of the Q1, Q2 switches and the Q3, Q4 cascode devicesare given by:V_(CE)(Q1,Q2)=f₁(VCC,Imod,Ibias)V_(CE)(Q3,Q4)=f₂(VCC,Imod,Ibias)

In order to be able to use the cascoded output switch architecture whileoperating at a low supply voltage, the modulation current is generatedby a simple resistor Rmod (most prior art use transistor based currentsources that need more headroom and thus a higher supply voltage). Thetail resistor Rmod is required to have as high a value as possible toassure a low variation of the modulation current with the variation ofthe voltage at the common emitter point of Q1 and Q2 during theswitching process. The maximum value of the Rmod resistor is limited bythe given supply voltage and the required headroom on the switch andcascode devices. Usually Rmod is only few Ohms, which leads tosignificant modulation current change during the switching process. Toreduce this parasitic variation of the modulation current, an additionalinductance (Lmod, FIG. 12) is connected in series with the tailresistor. This increases the AC impedance of the tail branch, whilekeeping the same DC impedance (assuming that the inductance hasnegligible series resistance). The tail inductance needs to withstandlarge currents (up to 120 mA). If on-chip inductances are used, theyneed to use very wide metal traces that lead to significant parasiticcapacitances (Cpar) that short-out the inductance at high frequenciesand thus cancel the beneficial effect of AC impedance increasing.

The present design uses a long bond wire as the tail inductance. Theparasitic series resistance is low and does not affect the headroom,while the AC impedance at the data-rate frequency is several timeshigher than the tail resistance (Rmod), even with a 1–2 nH inductance.

Thus, the modulation current is set by a separate common mode DCfeedback that imposes a voltage at the base of the transistor Q9, andthrough the base-emitter voltages of Q9, Q5/Q6 and Q1/Q2, regulates thevoltage on the tail resistor (Vmod) that gives the value of themodulation current (Imod=Vmod/Rmod). The Imod current value is set bythe voltage Vmod given by the resistor divider Rmod1 and Rmod2 thatprovides the reference voltage for the non-inverting input of the OAmodoperational amplifier. The inverting input of OAmod is connected to theRmod tail resistor and determines thatVmod=(Vmodcontrol)*Rmod2/(Rmod1+Rmod2).

In laser diode/EAM drivers, the modulation current is required to varyover very wide ranges (e.g. from 10 mA up to 120 mA). Keeping therise/fall time under a maximum specified value while achieving anovershoot/undershoot no higher than a given value requires that both thevoltage swing and the current level in the predriver be dependent on themodulation current. The optimal switching at the output (low rise/falltime and small overshoot) is achieved if the differential drivingvoltage swing is:Vswing=6V _(T) +Imod*Rdegen

Where: Rdegen is the total emitter degeneration resistance (either theintrinsic device emitter resistance if no external degeneration isprovided, or the sum of the intrinsic emitter resistance and theexternal emitter degeneration if present), andVT=kT/q(the thermal voltage)

The thermal voltage (VT=kT/q) is dependent on the absolute temperatureof the driver die. At higher temperatures, a higher voltage swing isrequired. The optimal performance of the output switch is obtained ifthe driving swing has embedded therein the 6 VT temperature dependence.Also the driving swing needs to have a modulation current dependence inorder to balance the voltage drop across the total emitter degenerationresistance (Imod*Rdegen). The driving swing is generated by thecollector resistor (Rc) and the tail current at the predriverdifferential pair (Q10, Q11). The voltage swing required by theforegoing equation is provided by using two tail currents for thepredriver differential pair: a PTAT current generated through the use ofa PTAT Bandgap Reference that gives the 6VT voltage drop across thecollector resistance (Rc), and a modulation current dependent currentsource Imod*=n*Imod that gives the Imod*Rdegen voltage drop across thecollector resistance. In the circuit of FIG. 12, the modulation currentdependent current source is set through the Imod control, whereas in thecircuit of FIGS. 13 and 14, it is set together with the modulationcurrent through the Vmod control, as previously explained.

The output switch devices Q1 and Q2 are operated very close to theirtransition frequency (f_(T)/5 . . . f_(T)/3), which implies a low basecurrent gain (β=3 . . . 5). In order to supply the high AC base currentrequired by the output switch, the emitter follower of the predriverneeds to also run at high current levels.

Most prior art uses a standard emitter follower with a tail currentvariable with the modulation current (Imod). This assures a lowvariation of the turn-off time for the entire range of modulationcurrents. The output switch transistors need significantly lower pull-upcurrent at turn-on than the pull-down current at turn-off(Ipull_up=Ipull_down/3 . . . 5). If the emitter follower current is thesame at turn-on and turn-off and given by the turn-off time condition,excessive overshoot appears at the turn-on, degrading the deterministicjitter.

Some of the prior art laser diode/EAM drivers use a dynamic emitterfollower that has a different current value when pulling-up (duringturn-on) versus when pulling-down (during turn-off). One way toimplement a dynamic emitter follower is to use one constant currentsource (Ifix) for each emitter follower and a differential pair drivenin opposition of phase with respect to the emitter follower that willprovide all its tail current (Idif) to the emitter follower that ispulling down (this is true if the delay of the helper differential pairis lower than the main signal path delay). The resulting pull-up andpull-down currents are given by:Ipull_up=IfixIpull_down=Ifix+Idif

The current at the differential pair (Idif) can be made dependent on themodulation current (Imod) and thus a variable driving capability of theemitter follower that pulls-down is assured as the modulation currentchanges, leading to a low variation of the rise/fall time over a widecurrent range. The drawback of this architecture is that the turn-oncurrent is constant for all the modulation current range, leading toexcessive overshoot at turn-on for low modulation current values.

The present invention adds a modulation current dependent pull-upcurrent to the dynamic emitter follower, assuring very low variation ofthe overshoot for the entire range of modulation currents. Themodulation current independent portion of the pull-up and pull-downcurrents are provided by a PTAT current source, assuring an optimalcompromise between rise/fall time and overshoot/undershoot for a widetemperature range. The circuits of FIGS. 12, 13 and 14 provide an Imoddependent current for both turn-on and turn-off currents, optimizing theswitching over a wide modulation current range. The PTAT and Imoddependencies of both voltage swing and current levels in the predriverlead to a much smaller variation (with respect to the prior art) of therise/fall time and overshoot/undershoot of the output waveform (currentor voltage) over a wide temperature and modulation current range. Thusless overshoot compensation and lower margins can be used when designingthe optical transmitter system.

While the present invention has been illustrated using silicon-bipolardevices, the invention applies equally to heterojunction laser/EAMdrivers and to MOS or HEMPT drivers.

In summary, the present invention proposes a cascaded output switcharchitecture that minimizes the Miller effect by providing a lowimpedance at the collectors of the output switch. Also the inventionminimizes the output capacitance of the driver, increasing the frequencyof the ringing and thus assuring an easier filtering. To keep therequired headroom on the output switch while operating at the samesupply voltages as the standard non-cascoded output switch, the tailcurrent of the output switch (modulation current) is generated by asimple tail resistor. A common-mode feedback loop is added to regulatethe voltage at the common emitter point and thus to impose the value ofmodulation current (Imod). The headroom of both the switch and thecascode device are optimized by a specially designed cascode biasgenerator that includes supply voltage, modulation current (Imod) andbias current (Ibias) dependencies.

In the case of EAM drivers, the cascoded switch architecture allowson-chip summation of the bias current directly at the emitter of theactive cascode device, thus reducing the parasitic modulation of thebias current by the high amplitude output voltage, and also reducing theoutput capacitance of the driver.

The cascoded switch architecture minimizes both the output capacitanceand capacitive loading on the predriver leading to a low deterministicjitter and a low overshoot of the output waveform while achieving a highedge speed.

The present invention uses a new temperature compensation technique thatconsists of including a positive temperature coefficient in both thedriving voltage swing and the pull-up/pull-down currents of thepre-driver in order to assure a very low variation of the edge speed andto minimize the peaking variation over temperature. In presentinvention, PTAT current sources are used to automatically adjust thevoltage swing and the predriver current level with the actual drivertemperature. Also Imod dependence of voltage swing and predriver currentlevels at both turn-on and turn-off perform an automatic adjustment thatkeep constant the rise/fall time and the overshoot/undershoot over awide range of modulation current.

The present invention also uses an alternative way of achieving theon-chip summation of the bias and modulation current, by doing it at alow impedance node and thus minimizing the edge speed penalty.

While certain preferred embodiments of the present invention have beendisclosed and described herein, it will be understood by those skilledin the art that various changes in form and detail may be made thereinwithout departing from the spirit and scope of the invention. Similarly,the various aspects of the present invention may be advantageouslypracticed by incorporating all features or various sub-combinations offeatures in any specific LD/EAM design.

1. A laser diode/electro-absorption-modulator (LD/EAM) drivercomprising: a cascoded output switch having a pair of output devices anda pair of cascode devices; a resistor providing tail current to theoutput devices; a predriver circuit receiving an input signal andcontrolling the output devices; a feedback circuit coupled to theresistor to control a modulation current of the output devices bycontrol of bias on the predriver circuit; a common mode feedback circuitproviding modulation dependent currents for the predriver circuit; and,a cascode bias circuit coupled to bias the cascode devices to a biasvoltage responsive to a power supply voltage, the output bias currentand the modulation current.
 2. The LD/EAM driver of claim 1 furthercomprised of an output bias circuit providing for on-chip summation ofthe modulation and an output bias current at a low impedance node of theactive cascode device.
 3. The LD/EAM driver of claim 1 further comprisedof a PTAT bandgap reference circuit to generate biasing currents withpositive temperature coefficients for the predriver circuit.
 4. TheLD/EAM driver of claim 3 wherein the modulation current is externallyadjustable.
 5. The LD/EAM driver of claim 1 wherein the modulationcurrent is externally adjustable.
 6. The LD/EAM driver of claim 1wherein the LD/EAM driver is an integrated circuit and the predriverbias current control and the modulation current are externallyadjustable.
 7. The LD/EAM driver of claim 1 wherein the LD/EAM driver isan integrated circuit and the predriver bias current control and themodulation current are externally adjustable by a single externaladjustment.
 8. The LD/EAM driver of claim 1 wherein the LD/EAM driver isan integrated circuit and the predriver bias current control and themodulation current are independently externally adjustable.
 9. TheLD/EAM driver of claim 1 further comprised of a pulldown variancecircuit coupled to the predriver, the pulldown variance circuit causinga turnoff current of the predriver to be larger than a turn-on currentof the predriver.
 10. The LD/EAM driver of claim 9 further comprised ofa PTAT bandgap reference circuit to generate biasing currents withpositive temperature coefficients for the predriver circuit.
 11. TheLD/EAM driver of claim 10 wherein the pulldown variance circuit isresponsive to the output of the bandgap reference.
 12. A laserdiode/electro-absorption-modulator (LD/EAM) driver comprising: acascoded output switch having a pair of output devices and a pair ofcascode devices; a resistor providing tail current to the outputdevices; a predriver circuit receiving an input signal and controllingthe output devices; a feedback circuit coupled to the resistor tocontrol a modulation current of the output devices by control of bias onthe predriver circuit; a common mode feedback circuit providingmodulation dependent currents for the predriver; and, a pulldownvariance circuit coupled to the predriver, the pulldown variance circuitcausing a turnoff current of the predriver to be larger than a turn-oncurrent of the predriver.
 13. The LD/EAM driver of claim 12 furthercomprised of a PTAT bandgap reference circuit to generate biasingcurrents with positive temperature coefficients for the predrivercircuit.
 14. The LD/EAM driver of claim 13 wherein the pulldown variancecircuit is responsive to the output of the bandgap reference.
 15. Alaser diode/electro-absorption-modulator (LD/EAM) driver comprising: acascoded output switch having a pair of output devices and a pair ofcascode devices; a resistor providing tail current to the outputdevices; a predriver circuit receiving an input signal and controllingthe output devices; a feedback circuit coupled to the resistor tocontrol a modulation current of the output devices by control of bias onthe predriver circuit; a common mode feedback circuit providingmodulation dependent currents for the predriver circuit; a cascode biascircuit coupled to bias the cascode devices to a bias voltage responsiveto a power supply voltage, an output bias current and the modulationcurrent; and, a pulldown variance circuit coupled to the predriver, thepulldown variance circuit causing a turnoff current of the predriver tobe larger than a turn-on current of the predriver.
 16. The LD/EAM driverof claim 15 further comprised of a PTAT bandgap reference circuit togenerate biasing currents with positive temperature coefficients for thepredriver circuit.
 17. The LD/EAM driver of claim 16 wherein thepulldown variance circuit is responsive to the output of the bandgapreference.
 18. The LD/EAM driver of claim 17 wherein the pulldownvariance circuit is responsive to the output of the bandgap reference.19. A laser diode/electro-absorption-modulator (LD/EAM) drivercomprising: a cascoded output switch having a pair of output devices anda pair of cascode devices; a resistor providing tail current to theoutput devices; a predriver circuit receiving an input signal andcontrolling the output devices; a feedback circuit coupled to theresistor to control a modulation current of the output devices bycontrol of bias on the predriver circuit; a common mode feedback circuitproviding modulation dependent currents for the predriver circuit; acascode bias circuit coupled to bias the cascode devices to a biasvoltage responsive to a power supply voltage, an output bias current andthe modulation current; a PTAT bandgap reference circuit to generatebiasing currents with positive temperature coefficients for thepredriver circuit; and, a pulldown variance circuit coupled to thepredriver, the pulldown variance circuit causing a turnoff current ofthe predriver to be larger than a turn-on current of the predriver. 20.The LID/EAM driver of claim 19 wherein the modulation current isexternally adjustable.
 21. The LD/EAM driver of claim 19 wherein theLD/EAM driver is an integrated circuit and the predriver bias currentcontrol and the modulation current are externally adjustable.
 22. TheLD/EAM driver of claim 19 wherein the LD/EAM driver is an integratedcircuit and the predriver bias current control and the modulationcurrent are externally adjustable by a single external adjustment. 23.The LD/EAM driver of claim 19 wherein the LD/EAM driver is an integratedcircuit and the predriver bias current control and the modulationcurrent are independently externally adjustable.
 24. The LD/EAM driverof claim 23 wherein the pulldown variance circuit is responsive to theoutput of the bandgap reference.